The invention relates generally to the technique of taking discrete samples from a signal, like a received signal at a radio frequency or intermediate frequency in a receiver. Especially the invention relates to reducing the inherent aliasing of noise in the sampling process.
A transmitted radio signal contains some information modulated onto a radio frequency carrier. A multitude of radio receiver architectures are known for receiving the transmitted radio frequency signal and downconverting the received signal into baseband where the information content of the signal may be reconstructed. It is common to use a superheterodyne receiver which converts the received signal first into an intermediate frequency (IF), where some amplification and filtering is performed, and to apply a second downconversion from IF to baseband. Previously a direct conversion receiver has been proposed for reducing power consumption and decreasing the space taken by the components of the radio receiver.
FIG. 1 illustrates a known direct conversion receiver 100, where a radio frequency signal picked up by the antenna 101 is filtered in a band pass filter 102, which is also called a preselection filter, and amplified in a Low Noise Amplifier (LNA) or preamplifier 103 before mixing it into baseband simultaneously in two parallel mixers 104 and 105. The mixers share a common local oscillator (LO) 106 but the LO signal is phase shifted by xcfx80/2 radians in a phase shifter 107 before feeding it into one of the mixers to produce a pair of mutually phase shifted mixing results called the I and Q signals. The mixing result from each mixer is filtered in a low pass filter 108 or 109 before converting it into a stream of digital samples in an analogue to digital (A/D) converter 110 or 111.
The drawbacks of the arrangement of FIG. 1 are its inferior sensitivity compared to that of a corresponding superheterodyne receiver and LO leakage owing to the fact that the LO frequency is situated in the operational frequency band of the Low Noise Amplifier 103 and the preselection filter 102.
FIG. 2 illustrates an alternative approach 200 to downconversion into baseband, known as subsampling. A radio frequency signal or an intermediate frequency signal is conducted along an input line 201 through a bandpass filter 202 to the input of a switch 203. The output of the switch is coupled to the input of an amplifier 204 and to a capacitor 205, the other end of the latter being connected to a reference potential, which is usually ground potential. The control signal 206 that controls the state of the switch 203 is a square wave coming from a local oscillator at a frequency which is either an integral multiple or a subharmonic of the radio frequency being downconverted. The frequency of the control signal is called the clock frequency or sampling frequency. The amplifier 204 acts as an output buffer. The arrangement of switch 203 and capacitor 205 is generally called an xe2x80x9cswitched-capacitor samplerxe2x80x9d and will be denoted as an xe2x80x9cSC samplerxe2x80x9d from here on. The buffered output of an SC sampler on line 207 is a baseband signal. During the time interval when switch 203 is closed the SC sampler is said to be in tracking mode and during the time interval when switch 203 is open the SC sampler is said to be in hold mode.
One of the problems in an arrangement according to FIG. 2 is the limited speed of the amplifier 204. It is commonplace to use a CMOS amplifier because of the advantageous features inherent to CMOS technology. However, a known CMOS amplifier (when driven at a reasonable power level) is so slow to react to the changes in its input that the clock frequency of the arrangement must remain below 100 MHz. The SC sampler itself (the combination of a controllable switch and a capacitor) could operate at a much higher clock frequency.
The consequences of a relatively low sampling frequency are seen in the noise figure of the arrangement. The total noise figure NFTOT of a receiver front-end comprising an LNA and a subsampling arrangement according to FIG. 2 can be expressed as                                           NF            TOT                    =                                                                      B                  N                                                  f                  N                                            ⁢                              NF                LNA                                      +                                                            NF                  mix                                -                1                                            G                LNA                                                    ,                            (        1        )            
where Bn is the equivalent noise bandwidth at the LNA output, fn is the Nyquist frequency of the sampler, NFLNA is the noise figure of the LNA, NFmix is the noise figure of the subsampling arrangement (also known as the subsampling mixer) and GLNA is the gain factor of the LNA. The first term in, (1) shows that to minimize the noise figure it is advantageous to limit the bandwidth before the sampler and maximize the sampling frequency. However, in an integrated circuit it may often be impossible or expensive to reduce Bn sufficiently to prevent noise aliasing. In the second term of (1) the factor NFmix depends on the sampling frequency according to the formula                                           NF            mix                    =                      1            +                          1                              4                ⁢                                  C                  h                                ⁢                                  R                  S                                ⁢                                  f                  N                                                                    ,                            (        2        )            
where Ch is the capacitance of the sampling capacitor and RS is the source resistance of the subsampling mixer. In the derivation of (2) the usual assumption was made that the total noise power may be expressed as kT/Ch. Although NFmix is typically high for subsampling mixers, it may be noted from equation (1) that it is divided by the preceding gain and therefore presents no fundamental limitation.
When a very fast slewing signal is sampled, the dynamic range of the subsampling mixer is degraded by timing uncertainty in the clock frequency. It can be shown that in the sampling-based conversion of a signal with frequency f, a jitter referenced to as tj limits the Signal-to-Noise Ratio (SNR) to                               SNR          =                      10            ·                                          log                10                            ⁡                              [                                  OSR                                                                                    (                                                  2                          ⁢                          π                          ⁢                                                      xe2x80x83                                                    ⁢                          f                                                )                                            2                                        ⁢                                          t                      j                      2                                                                      ]                                                    ,                            (        3        )            
where the oversampling ratio OSR is defined as the ratio between the signal bandwidth and the sampler Nyquist frequency. Oversampling data converters are insensitive to jitter because their OSR is high and the signal frequency f is respectively low. On the contrary, a subsampling mixer always sees a relatively high input frequency f; increasing the sampling rate nevertheless reduces the effect of timing jitter because OSR is thus increased.
It is an object of the present invention to bring forward a method and a circuit for downconverting an oscillating signal, the noise characteristics of the circuit being better than those of prior art solutions. A further object of the invention is that the presented method would be equally applicable to direct downconversion from radio frequency and to downconversion from intermediate frequency. A still further object if the invention is that the resulting circuit is efficient in terms of pwer consumption.
The objects of the invention are achieved by placing at least two SC samplers in parallel and using an amplifier to simultaneously buffer the outputs of the parallel SC samplers.
The electrical circuit according to the invention has an input and an output and is meant for
sampling an input signal coupled to the input having a certain input frequency and
converting the input signal into a certain output frequency at the output, the output frequency being lower than the input frequency. It is characterised in that it comprises
a first sampler circuit coupled to the input,
a second sampler circuit coupled to the input,
a buffering component coupled to the output and
buffer switching means arranged to respond to a buffering command by coupling said first sampler circuit and said second sampler circuit to said buffering component.
The invention also concerns a receiver which is characterised in that it comprises in a sampling downconverter block:
a first sampler circuit coupled to the input of the sampling downconverter block,
a second sampler circuit coupled to the input of the sampling downconverter block,
a buffering component coupled to the output of the sampling downconverter block and
buffer switching means arranged to respond to a buffering command by coupling said first sampler circuit and said second sampler circuit to said buffering component.
Additionally the invention concerns a method for sampling and downconverting a signal. The method according to the invention is characterised in that it comprises the steps of
taking a first sample from the input signal during a first sampling time interval and holding said first sample,
taking a second sample from the input signal during a second sampling time interval, which is later in time than said first sampling time interval, and holding aid second sample,
taking a combination of said held first sample and said held second sample and
buffering said combination in an output buffer.
As mentioned earlier, an SC sampler is capable of operating at a much higher clock frequency than what a conventional buffering CMOS amplifier tolerates. An obvious solution to this incongruity would be to use a more elaborate amplifier or some other trick to enhance the frquency tolerance of the amplifier. However, this would easily result in large amounts of power being used in the amplifier, giving rise to temperature problems and untolerably high consumption of energy. The latter is a very serious drawback in battery-powered modern mobile radio apparatuses like mobile telephones.
The invention approaches the problem from a totally different viewpoint. If several SC samplers are connected in parallel, a relatively high first clock frequencyxe2x80x94for example in the order of several hundreds of MHzxe2x80x94may be used to Digger the sampling event in each SC sampler at a different moment of time. A time period during which exactly one clock pulse will be given to each one of the parallel SC samplers is called a sampling cycle. A second clock frequency is a fraction of the first clock frequency and it is used to couple the parallel SC samplers to a common output buffer (advantageously a buffering amplifier) simultaneously, once during each sampling cycle. The second clock frequency is not higher than what the buffering amplifier can tolerate.
The effect of the invention on the noise characteristics of a subsampling receiver front-end is significant, as can be seen from the formulae presented previously. The sampling frequency that appears in (1)-(3) through the Nyquist frequency fn is the first clock frequency defined above, which is several times higher than the second clock frequency that is limited by the capabilities of the amplifier or by the power budget of the system. As the Nyquist frequency fn gets higher, the first term in equation (1) (the term that dominates the noise figure) gets lower and the maximun Signal-to-Noise Ratio SNR limited by the jitter gets higher.
A very advantageous further application of the invention may be presented, in which the polarity of a number of samples is additionally inverted before coupling the samples from the parallel SC samplers to the common buffering amplifier. At the time of filing this patent application it is seen that in the most advantageous embodiment of this further application of the invention, every other sample is inverted. The inversion is especially simple to implement if the signal to be sampled is differential, wherein an inversion corresponds to simply cross-connecting a pair of signal wires. This further application of the invention is applicable e.g. to produce a quasi-direct conversion from radio frequency to baseband, in which the received signal is first converted to IF by sampling and immediately moved to baseband by inverting every other sample.
A buffering amplifier is used throughout this patent application as an example of an output component; however, the parallel SC sampler branches could as well be connected to some other kind of output component, like an A/D converter or a filter or any other signal processing block accepting an SC input. An amplifier is generally an advantageous output component, because it buffers the signal as a voltage and not as a charge, whereby mismatch between the capacitances is not a problem. To avoid confusing multiple definitions for practically the same thing, the component to which the parallel SC sampler branches are connected is generally called a buffering component. Similarly the operation of reading the outputs of the parallel SC sampler branches into the output component is called simply buffering.